As is known in the art, phased array antennas include a plurality of antenna elements spaced apart from each other by known distances coupled through a plurality of phase shifter circuits to either or both of a transmitter or receiver.
As is also known, phased array antenna systems produce a beam of radio frequency energy (RF) and direct such beam along a selected direction by controlling the phase (via the phase shifter circuitry) of the RF energy passing between the transmitter or receiver and the array of antenna elements. In an electronically scanned phased array, the phase of the phase shifter circuits (and thus the beam direction) is selected by sending a control signal or word to each of the phase shifter sections. The control word is typically a digital signal representative of a desired phase shift, as well as a desired attenuation level and other control data.
Phased array antennas are often used in both defense and commercial electronic systems. For example, active electronically scanned arrays (AESAs) are in demand for a wide range of defense and commercial electronic systems such as radar surveillance, terrestrial and satellite communications, mobile telephony, navigation, identification, and electronic counter measures. Such systems are often used in radar for land base, ship and airborne radar systems and satellite communications systems. Thus, the systems are often deployed on a single structure such as a ship, aircraft, missile system, missile platform, satellite, or building where a limited amount of space is available.
AESAs offer numerous performance benefits over passive scanned arrays as well as mechanically steered apertures. However, the costs associated with deploying AESAs can limit their use to specialized military systems. An order of magnitude reduction in array cost could enable widespread AESA insertion into military and commercial systems for radar, communication, and electronic warfare (EW) applications. The performance and reliability benefits of AESA architectures could extend to a variety of platforms, including ships, aircraft, satellites, missiles, and submarines. Increasing modularity, reducing fabrication costs and increasing the quantity of components being manufactured can drive down the unit costs of the components and thus the cost of the AESAs.
With the desire to reduce the cost of array antennas, and in particular the cost of array antennas having relatively large apertures, it has become common to develop the antenna aperture as an array of active sub-array apertures. These sub-arrays typically have their own internal RF power dividers, driver amplifiers, time delay units, logic distribution networks, DC power distribution networks, DC/DC converters, and accessible ports for RF, logic, DC power, and thermal management interfaces. It would desirable if each of the sub-arrays could be manufactured the same and be used interchangeably in the fabrication of the complete array. But when the aperture is formed from sub-arrays, it has, heretofore, lacked flexibility because the RF distribution networks required for receive beam formation and exciter output distribution are hardwired into the aperture backplane and are thus position dependent. Thus, typical AESA apertures are not configured such that the sub-arrays are interchangeable.
To further complicate the problem, a tracking radar having a highly directive antenna pattern (narrow main beam) seeks to keep the antenna electrical boresite aligned with a target of interest. The method typically used to track targets is monopulse beamforming where the angular location of a target is obtained by comparison of signals received simultaneously via three antenna patterns (called the “sum pattern”, “elevation monopulse pattern” and “azimuth monopulse pattern”; typically the monopulse patterns are referred to more simply as the “El pattern” and “Az pattern”).
Two conventional approaches for AESA monopulse are: (1) analog beamforming; and (2) combined analog-digital beamforming. In the analog beamforming approach, an analog RF feed network combines multiple AESA transmit/receive (T/R) channels into sub-arrays; then sub-arrays are combined via unique RF feed networks which couple and weight sub-array RF receive signals to produce an array-level monopulse pattern in elevation and azimuth angle.
In the combined analog-digital beamforming approach, an analog RF feed network combines multiple AESA T/R channels into sub-arrays in the same manner as the all-analog approach. Analog to Digital (A/D) converters at each sub-array produce digital signals that are then combined to form the final array-level monopulse pattern in elevation and azimuth angle.
Thus, elevation and azimuth monopulse patterns can be generated with analog beamforming techniques, digital beamforming techniques, or a combination of both analog and digital beamforming.
To ensure accurate beamsteering in a phased array, it is necessary to calibrate the array by determining path length differences in each RF signal path leading to an antenna element and adjusting for such differences.
In a planar AESA, three-channel monopulse (receive sum channel, receive delta elevation channel and receive delta azimuth channel) calibration is typically accomplished using a time-intensive process which uses waveguide-probe measurements in a planar near-field range. Not only are conventional calibration techniques time-consuming, such approaches sacrifice pattern accuracy because calibration is performed with only one transmit/receive (T/R) channel on at a time. That is, the receive sum, receive delta elevation and receive delta azimuth channels are all calibrated in the sub-array near-field; therefore requiring that the waveguide probe move from T/R channel to T/R channel which is time-consuming and thus expensive to perform on a large AESA. This leads to a trade-off in near-field calibration time versus calibration accuracy. As described herein, calibration accuracy relates to the receive sum channel, receive delta elevation channel and receive delta azimuth channel antenna requirements for 1) sidelobe level, 2) beam pointing accuracy and 3) delta pattern null depth (a significant driver of monopulse slope).
As noted above, the importance of monopulse beamforming in radar performance is well-known.
Elevation and azimuth monopulse patterns can be generated with analog beamforming techniques or digital beamforming (prior to receiver matched filter processing) technique. Alternatively a combination of both analog and digital beamforming can be used. A brief description of each monopulse beamforming approach is given below.
When analog beamforming is used, the traditional approach is to design and fabricate a unique corporate RF feed network for each sub-array. However, any change to the AESA active aperture dimension, sub-array dimension or sidelobe performance specification requires a re-design of the beamforming network. Typically, a planar near-field scanner is used for three-channel monopulse calibration where a sequential set of measurements are made with one T/R channel commanded on at a time.
When all-digital beamforming is used, digital beamforming at the T/R element level provides design invariance to a change in AESA active aperture dimension, sub-array dimension or sidelobe performance. A planar near-field scanner could be used for calibration but the calibration needs to be performed only once (per frequency) since the RF signal is immediately digitized at the T/R channel level. Presently, digital beamforming at the T/R element level above L-Band is cost prohibitive.
Therefore, it is highly desired to have a calibration technique that reduces antenna calibration time while maximizing AESA monopulse pattern performance.
An AESA sub-array typically has dozens to hundreds of T/R channels. Calibration of the monopulse channels using conventional approaches is, therefore, a compromise between pattern accuracy (e.g., monopulse null/slope, sidelobes, beam-pointing accuracy) and near-field test time.
With respect to monopulse pattern accuracy, to achieve the highest accuracy for the receive delta elevation channel and receive delta azimuth channel, one measures the RF signal path through the receive delta elevation channel (for delta elevation channel calibration) and receive delta azimuth channel (for delta azimuth channel calibration). This must be done for all receive sum T/R channels in the panel (or sub-array). Then for each of the above-described RF measurements, all combinations of digital attenuation and digital phase state are measured through the receive delta elevation channel and receive delta azimuth channel respectively. Measuring all T/R channel states minimizes amplitude and phase errors, but it is also the most time-consuming.
With respect to test time, to achieve a satisfactory AESA calibration test time for the receive delta elevation channel and receive delta azimuth channel, one measures the RF path signal through the receive delta elevation channel (and receive delta azimuth channel) for only one (or several) receive sum T/R channels. For each of the above-described RF measurements, several combinations of digital attenuation and digital phase state are measured through the receive delta elevation channel (and receive delta azimuth channel). Measuring several T/R channel states, instead of all states, results in higher amplitude and phase errors, but it is also reduces test time.
Long near-field test times add considerable labor costs to an AESA; but, AESA monopulse tracking accuracy is directly dependent on the monopulse null/slope, beam-pointing accuracy and relatively low sidelobes (compared to the receive sum channel sidelobes), which require a more complete set of near-field measurements that reduce residual errors in amplitude and phase due to the digital attenuator and phase shifter at the T/R channel level.
It would, therefore, be desirable to provide an AESA calibration technique that simultaneously reduces near-field calibration time and reduces (or ideally minimizes) errors in receive delta elevation and delta azimuth channels.